Radar front end with RF oscillator monitoring

ABSTRACT

An apparatus is described that, according to an exemplary embodiment, has an RF oscillator for generating an RF oscillator signal at a first frequency and a frequency divider having a division ratio that is fixed during operation. The frequency divider is supplied with the RF oscillator signal and is configured to provide an oscillator signal at a second frequency. The apparatus further has a monitor circuit, to which the oscillator signal at the second frequency is supplied and which is configured to measure the second frequency and to provide at least one digital value that is dependent on the second frequency of the oscillator signal. The at least one digital value is provided on a test contact.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/014,044 filed Jun. 21, 2018, which claims the benefit of GermanPatent Application No. 10 2017 113 730.0 filed Jun. 21, 2017, which areincorporated by reference as if fully set forth.

FIELD

The present disclosure relates to the field of radio-frequency (RF)circuits. Some exemplary embodiments relate to a radar front end havingan RF oscillator and a monitoring circuit for checking the operation ofthe RF oscillator.

BACKGROUND

Radio-frequency (RF) transmitters and receivers are found in amultiplicity of applications, particularly in the field of wirelesscommunication and radar sensors. In the automotive sector, there is anincreasing need for radar sensors, which are used in what are known asadaptive cruise control (ACC, or radar cruise control) systems. Suchsystems can automatically adapt the speed of an automobile so as to keepa safe distance from other automobiles traveling ahead (and from otherobjects and from pedestrians). Further applications in the automotivesector are e.g. blind spot detection, lane change assist and the like.

Modern radar systems use large-scale integrated RF circuits that cancombine all the core functions of an RF front end of a radar transceiverin a single package (single-chip radar transceiver), this frequentlybeing referred to as an MMIC (monolithic microwave integrated circuit).Such RF front ends usually contain, inter alia, a voltage-controlledoscillator (VCO) connected in a phase locked loop, power amplifiers(PA), directional couplers, mixers and analog-to-digital convertors(ADC), and also associated control circuit arrangements for controllingand monitoring the RF front end. Radar applications used in automobilesare subject to various standards relating to safety in road traffic, forexample, the functional safety standard ISO 26262 entitled “Roadvehicles—Functional safety”. In order to ensure the functional safety ofa radar sensor and/or to meet legal regulations, the RF front end shouldoperate with well-defined operating parameters.

It would be desirable to provide an integrated radar system havingimproved self-test or self-monitoring capabilities.

SUMMARY

The aforementioned object may be achieved by the apparatus according toone or more described embodiments. Various embodiments and furtherdevelopments are the subject of the claims.

An apparatus is described that, according to an exemplary embodiment,has an RF oscillator for generating an RF oscillator signal at a firstfrequency and a frequency divider having a division ratio that is fixedduring operation. The frequency divider is supplied with the RFoscillator signal and is configured to provide an oscillator signal at asecond frequency. The apparatus further has a monitor circuit, to whichthe oscillator signal at the second frequency is supplied and which isconfigured to measure the second frequency and provide at least onedigital value that is dependent on the second frequency of theoscillator signal. The at least one digital value is provided on atleast one test contact.

Moreover, a method is described that, according to an exemplaryembodiment, involves the following: actuating a voltage-controlled RFoscillator such that it generates an RF oscillator signal at a firstfrequency; generating an oscillator signal at a second frequency, fromthe first RF oscillator signal, by means of a frequency divider having adivision ratio that is fixed during operation; measuring the secondfrequency and providing at least one digital value that is dependent onthe second frequency of the oscillator signal; and providing the atleast one digital value on at least one test contact.

Furthermore, a monolithic microwave integrated circuit (MMIC) isdescribed. According to an exemplary embodiment, the MMIC has an RFoscillator, integrated in a semiconductor chip, for generating an RFoscillator signal at a first frequency. The semiconductor chip furtherincorporates a frequency divider having a division ratio that is fixedduring operation. The frequency divider is supplied with the RFoscillator signal and is configured to provide an oscillator signal at asecond frequency. The MMIC further has a monitor circuit, integrated inthe semiconductor chip, to which the oscillator signal at the secondfrequency is supplied and which is configured to measure the secondfrequency and provide at least one digital value that is dependent onthe second frequency of the oscillator signal. The at least one digitalvalue is provided on a test contact arranged on the semiconductor chip.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is explained in more detail below on the basis of theexamples depicted in the figures. The depictions are not necessarily toscale and the invention is not restricted only to the depicted aspects.Rather, emphasis is placed on depicting the principles on which theinvention is based. In the figures:

FIG. 1 shows a sketch to illustrate the functional principle of afrequency-modulated continuous wave radar system (FMCW) radar system fordistance and/or speed measurement.

FIG. 2 includes two timing diagrams to illustrate the frequencymodulation of the RF signal generated by the FMCW system.

FIG. 3 is a block diagram to illustrate the basic structure of an FMCWradar system.

FIG. 4 is a block diagram to illustrate an example of an analog RF frontend of the FMCW radar system from FIG. 3 .

FIG. 5 is a block diagram for illustrating an example of a phase lockedloop (PLL) for generating a frequency-modulated RF signal.

FIG. 6 illustrates an example of a phase locked loop having a monitorcircuit for monitoring and testing the voltage-controlled oscillator(VCO) contained in the phase locked loop.

FIG. 7 uses a timing diagram to illustrate a frequency ramp and theevaluated time intervals.

FIG. 8 uses timing diagrams to illustrate the frequency determination ofthe LO signal by means of counters.

FIG. 9 illustrates an exemplary implementation of the monitor circuitfrom FIG. 6 .

FIG. 10 is a flow chart illustrating an example of a method for testingthe VCO of an RF front end.

FIG. 11 is a flow chart illustrating an example of a method forautomatically tuning the VCO parameter k_(VCO).

FIG. 12 uses a timing diagram to illustrate a method for testing thelinearity of a frequency ramp.

DETAILED DESCRIPTION

The exemplary embodiments that now follow are described within thecontext of a radar receiver. The invention is not restricted to radarapplications, however, and can also be employed in other areas, forexample in RF transceivers of RF communication apparatuses. RF circuitsfrom a wide variety of areas of application can have voltage-controlledoscillators (VCOs) for generating RF signals. Instead of VCOs, it isalternatively also possible for digitally controlled oscillators (DCOs)to be used. The concepts that are now described can easily betransferred without hesitation to applications in which DCOs instead ofVCOs are used.

FIG. 1 illustrates the application of a frequency modulated continuouswave (FMCW) radar system as a sensor for measuring distances and speedsof objects, which are usually referred to as radar targets. In thepresent example, the radar apparatus 10 has separate transmission (TX)and reception (RX) antennas 5 and 6 (bistatic or pseudo-monostatic radarconfiguration). However, it should be noted that it is also possible fora single antenna to be used that simultaneously serves as a transmissionantenna and a reception antenna (monostatic radar configuration). Thetransmission antenna 5 radiates a continuous RF signal s_(RF)(t) that isfrequency-modulated, for example by means of a sawtooth signal(periodic, linear ramp signal). The radiated signal s_(RF)(t) isscattered back at the radar target T, and the backscattered (reflected)signal y_(RF)(t) is received by the reception antenna 6.

FIG. 2 illustrates the aforementioned frequency modulation of the signals_(RF)(t) by way of example. As depicted in FIG. 2 , the signals_(RF)(t) is composed of a set of “chirps”, i.e. signal s_(RF)(t)comprises a sequence of sinusoidal signal profiles (waveforms) at rising(up-chirp) or falling (down-chirp) frequency (see top diagram in FIG. 2). In the present example, the instantaneous frequency f(t) of a chirprises linearly at a starting frequency f_(START), beginning within atime period T_(RAMP), to a stopping frequency f_(STOP) (see bottomdiagram in FIG. 2 ). Such chirps are also referred to as a linearfrequency ramp. FIG. 2 depicts three identical linear frequency ramps.However, it should be noted that the parameters f_(START), f_(STOP),T_(RAMP) and the pause between the individual frequency ramps can vary.The frequency variation also does not necessarily have to be linear.Depending on the implementation, it is also possible for transmissionsignals with exponential (exponential chirps) or hyperbolic (hyperbolicchirps) frequency variation to be used.

FIG. 3 is a block diagram depicting a possible structure of a radarapparatus 1 (radar sensor) by way of example. Similar structures canalso be found e.g. in RF transceivers used in other applications, suchas e.g. wireless communication systems. Accordingly, at least onetransmission antenna 5 (TX antenna) and at least one reception antenna 6(RX antenna) are connected to an RF front end 10, which can contain allthose circuit components that are needed for the RF signal processing.These circuit components comprise, by way of example, a local oscillator(LO), RF power amplifiers, low noise amplifiers (LNA), directionalcouplers (e.g. rat-race couplers, circulators, etc.) and mixers fordown-converting the RF signals to baseband or an intermediate frequencyband (IF band). The RF front end 10 may—possibly together with furthercircuit components—be integrated in a monolithic microwave integratedcircuit (MMIC). The example depicted shows a bistatic (orpseudo-monostatic) radar system having separate RX and TX antennas. Inthe case of a monostatic radar system, a single antenna (or an antennaarray) would be used both for radiating and for receiving theelectromagnetic (radar) signals. In this case, a directional coupler(e.g. a circulator) can be used to separate the RF signals to beradiated to the radar channel from the RF signals (radar echoes)received from the radar channel.

In the case of a frequency-modulated continuous wave radar system (FMCWradar system), the RF signals radiated via the TX antenna 5 may be e.g.in the region of approximately 20 GHz and 81 GHz (e.g. 77 GHz in someapplications). As mentioned, the RF signal received by the RX antenna 6comprises the radar echoes, i.e. those signal components that arescattered back at what are known as the radar targets. The received RFsignal y_(RF)(t) is e.g. down-converted to baseband and processedfurther in baseband by means of analog signal processing (see FIG. 3 ,analog baseband signal processing chain 20). The cited analog signalprocessing essentially comprises filtering and possibly amplification ofthe baseband signal. The baseband signal is finally digitized (see FIG.3 , analog-to-digital convertor 30) and processed further in the digitaldomain. The digital signal processing chain may be realized at least inpart as software, which is executed on a processor (see FIG. 3 , DSP40). The entire system is normally controlled by means of a systemcontroller 50, which may likewise be implemented at least in part assoftware that can be executed on a processor, such as e.g. amicrocontroller. The RF front end 10 and the analog baseband signalprocessing chain 20 (optionally also the analog-to-digital convertor 30)may be jointly integrated in a single MMIC (i.e. an RF semiconductorchip). Alternatively, the individual components may also be distributedover multiple integrated circuits.

FIG. 4 illustrates an exemplary implementation of the RF front end 10with a downstream baseband signal processing chain 20, these possiblybeing part of the radar sensor from FIG. 3 . It should be noted thatFIG. 4 depicts a simplified circuit diagram in order to show the basicstructure of the RF front end. Actual implementations, which can behighly dependent on the specific application, may naturally be morecomplex. The RF front end 10 comprises a local oscillator 101 (LO) thatgenerates an RF signal s_(LO)(t). The signal s_(LO)(t) may, as describedabove with reference to FIG. 3 , be frequency-modulated and is alsoreferred to as an LO signal. In radar applications, the LO signal isusually in the SHF (super high frequency, centimeter-wave) or in the EHF(extremely high frequency, millimeter-wave) band, e.g. in a range from76 GHz to 81 GHz in automotive applications.

The LO signal s_(LO)(t) is processed both in the transmission signalpath and in the received signal path. The transmission signal s_(RF)(t)(cf. FIG. 2 ) radiated by the TX antenna 5 is generated by amplifyingthe LO signal s_(LO)(t), for example by means of the RF power amplifier102. The output of the amplifier 102 may be coupled to the TX antenna 5(in the case of a bistatic or pseudo-monostatic radar configuration).The received signal y_(RF)(t) provided by the RX antenna 6 is suppliedto the RF port of the mixer 104. In the present example, the RF receivedsignals y_(RF)(t) (antenna signal) is preamplified by means of theamplifier 103 (gain g), and the mixer 104 is supplied with the amplifiedRF received signal g y_(RF)(t). The amplifier 103 may be e.g. an LNA.The reference port of the mixer 104 is supplied with the LO signals_(LO)(t), so that the mixer 104 down-converts the (preamplified) RFreceived signal y_(RF)(t) to baseband. The down-converted basebandsignal (mixer output signal) is denoted by y_(BB)(t). This basebandsignal y_(BB)(t) is initially processed further in analog fashion,wherein the analog baseband signal processing chain 20 essentially hasamplification (amplifier 22) and filtering (e.g. band pass filter 21),in order to reject undesirable side bands at image frequencies. Theresulting analog output signal that can be supplied to ananalog-to-digital convertor is denoted by y(t). Methods for digitalfurther processing of the output signal (digital radar signal) are knownper se (for example range doppler analysis) and are therefore notdiscussed further now.

In the present example, the mixer 104 down-converts the preamplified RFreceived signal g y_(RF)(t) (i.e. the amplified antenna signal) tobaseband. The mixing can take place in one stage (that is to say fromthe RF band directly to baseband) or via one or more intermediate stages(that is to say from the RF band to an intermediate frequency band andon to baseband). In view of the example shown in FIG. 4 , it becomesclear that the quality of a radar measurement is highly dependent on thequality or accuracy of the LO signal s_(LO)(t). Depending on theapplication, radar sensors need to satisfy particular standards, forexample the functional safety standard ISO 26262. To ensure thefunctional safety of a radar sensor and/or to meet legal regulations,the RF front end should operate with well defined operating parameters.In the case of the RF front ends integrated in MMICs (e.g. single-chipradar), tolerances can mean that measures are needed during productionto check and calibrate particular parameters. During operation too, itwould be desirable to be able to monitor relevant parameters(monitoring) and, in the event of an inadmissible alteration of one ormore parameters, to signal an error in order to ensure that potentiallyunreliable measured values are detected as such.

As mentioned above, the quality or accuracy of the LO signal s_(LO)(t)is relevant to the quality of a radar measurement. The LO signals_(LO)(t) is normally generated by a voltage controlled oscillator(VCO). When monitoring the VCO, e.g. the absolute value and thelinearity of the VCO parameter k_(VCO) may be of interest. The VCOparameter k_(VCO) describes the ratio between control voltage V_(CTRL)of the VCO and the frequency f_(LO) of the output signal s_(LO)(t), thatis to say k_(VCO)=f_(LO)/V_(CTRL). A VCO is usually operated in a phaselocked loop (PLL). FIG. 5 shows a possible structure of a localoscillator (LO) with a VCO connected up in a PLL by way of example.

FIG. 5 shows a simplified circuit diagram that, by way of example, hasthe basic structure of a local oscillator comprising a PLL with a VCO.In the present example, the VCO 60 generates the RF oscillator signals_(LO)(t), which may be e.g. in the EHF band. The frequency f_(LO) ofthe RF oscillator signal s_(LO)(t) is dependent on the input voltageV_(CTRL) (control voltage) of the VCO 60. Since the frequency f_(LO) istoo high for direct further processing, the VCO 60 has a downstreamfrequency divider 61 having a constant division ratio 1/M. The divisor Mis an integer and may be e.g. 32. However, other values are alsopossible for M (e.g. 1, 2, 4, 6, 8, etc.). In the example of 1/M=1/32mentioned, an oscillator frequency f_(LO) of 80 GHz would be reduced to2.5 GHz. The divisor M is constant during operation, that is to say thatM does not change during operation. Nevertheless, M can be set to adesired value depending on the configuration of the RF front end. Thefrequency at the output of the frequency divider 61 is denoted byf_(LO)′ (f_(LO)′=f_(LO)/M); the reduced-frequency oscillator signal isdenoted by s_(LO)′(t).

The frequency divider 61 has a downstream multi-modulus divider 62 (MMD)that is configured to reduce the frequency f_(LO)′ of the signals_(LO)′(t) by a variable divisor N. The output signal of themulti-modulus divider 62 is denoted by s_(PLL)(t) and its frequency isdenoted by f_(PLL) (f_(LO)′/N=f_(PLL)). Continual variation of thedivision ratio 1/N of the multi-modulus divider 62 (e.g. by means of thesigma-delta modulator 63) can effectively produce a rational divisor. Inthis case, a desired rational divisor R is e.g. modulated by means ofthe sigma-delta modulator 63. At the output of the sigma-delta modulator63, an updated integer divisor value N is generated for themulti-modulus divider 62 in each clock cycle. Effectively—on average—arational divisor N is obtained. Such frequency divider circuits (MMD andmodulator) are also referred to as fractional N dividers. Thesigma-delta modulator can have e.g. a MASH (multi-stage noise shaping)structure.

The output signal s_(PLL)(t) of the multi-modulus divider 62 and areference signal s_(REF)(t) (frequency f_(REF)) are supplied to a phasedetector (PD) or phase-frequency detector (PFD) 64 configured to comparethe phases (or phases and frequencies) of the signals s_(PLL)(t) ands_(REF)(t). The output signal V_(CP) of the phase-frequency detector 64is dependent on the detected phase and/or frequency difference. Usually,the output stage of a phase-frequency detector comprises a charge pump.Various implementations of phase detectors and phase-frequency detectorsare known per se, however, and are not discussed further now. The outputsignal V_(CP) of the phase-frequency detector 64 is supplied to what isknown as the loop filter 65 (LF). This loop filter 65 essentiallydetermines the bandwidth of the PLL and, at its output, provides thecontrol voltage V_(CTRL) for the VCO 65, which closes the control loop.In a steady state, the phases of the signals s_(PLL)(t) and S_(REF)(t)are “locked” and the phases of the signals s_(PLL)(t) and s_(LO)(t) arein sync with the phase of the reference signal s_(REF)(t). The referencesignal s_(REF)(t) can be generated by means of a quartz oscillator orcan be generated based on a quartz oscillator signal (e.g. by means offrequency multiplication or frequency division, see e.g. FIG. 6 ), forexample.

FIG. 6 shows a further example of a local oscillator with a VCO 60arranged in a phase locked loop. The example is essentially the same asthe previous example from FIG. 5 , but has a few additional components.The reference signal s_(REF)(t) is generated in the present example byvirtue of the oscillator signal s_(CLK1)(t) of a reference oscillator 70(e.g. a crystal oscillator) being supplied to a frequency multiplier 71.The output signal of the frequency multiplier 71 is the reference signals_(REF)(t). The frequency multiplier factor of the frequency multiplier71 is 4 in the present example. In other exemplary embodiments, however,larger or smaller factors can be used. Moreover, FIG. 6 depicts theclock generator 72, which takes the PLL clock signal s_(PLL) (outputsignal of the MMD 62) as a basis for generating a clock signal for thesigma-delta modulator 63. In addition to the example from FIG. 5 , thephase locked loop shown in FIG. 6 also has a digital-to-analog convertor(DAC) 73, which is connected between the output of the ramp generator 70and the output of the loop filter 65. The DAC 73 can be used to directlyinfluence the control voltage V_(CTRL) for the VCO 60, as a result ofwhich it is possible for very large sudden frequency changes to be dealtwith. Otherwise, the example from FIG. 6 is the same as the previousexample from FIG. 5 , and reference is made to the explanations above.

The frequency modulation of the transmission signal s_(RF)(t), which isconsistent with the amplified LO signal s_(LO)(t), depicted by way ofexample in FIG. 2 can be accomplished by means of what is known as aramp generator 70 (RMP, see FIG. 5 ), for example, which is configuredto generate sequences of divisors that are supplied to the fractional Ndivider (MMD 62 and modulator 62). Variation of the (effectivelynon-integer) divisor N varies the frequency f_(LO) of the LO signals_(LO)(t) accordingly. The ramp generator 70 is configured to controlthe PLL, by varying the effective divisor of the MMD 62, such that theLO signal has the desired frequency modulation, i.e. the desiredstarting and stopping frequency, the desired chirp duration and thedesired timing (cf. FIG. 2 ). To monitor the LO signal s_(LO)(t), theoutput of the frequency divider 61 (constant division ratio 1/M) has amonitoring circuit 80 (MON) coupled to it. The LO signal s_(LO)(t) isthus monitored indirectly by virtue of the frequency f_(LO)′ of thefrequency-divided oscillator signal s_(LO)′(t) being analyzed. Themonitoring circuit is configured to generate a measured value for thefrequency f_(LO). The time control of the monitoring circuit 80 may bein sync with the time control of the ramp generator 70 (synchronizationsignal S_(TRIG)). In the present exemplary embodiments, the monitoringcircuit 80 is integrated in the same MMIC as the RF front end 10, bothsimplifying tests at the end of the production line (end-of-line tests)and allowing monitoring (self-tests) during the operation of the radarsensor. The operation of the monitoring circuit 80 is explained in moredetail below on the basis of examples.

The timing diagram from FIG. 7 shows the profile of the frequency f_(LO)and of the frequency f_(LO)′(t) during a frequency ramp by way ofexample. Between the times t₀ and t₁, a minimum oscillator frequency isset (f_(LO)=f_(MIN)); between the times t₁ and t₃, the oscillatorfrequency rises linearly to a maximum oscillator frequency(f_(LO)=f_(MAX)). After the time t₃, the oscillator frequency remainsconstant at f_(MAX). An exact frequency measurement may require asettling time T_(S) to be waited between the times t₁ and t₂ and also t₃and t₄ before the measurement is begun.

The timing diagrams from FIG. 8 illustrate the frequency measurementperformed by the monitoring circuit 80. A measurement is performedduring a measurement time interval T_(M) (time window) that issynchronized to a specific frequency ramp by means of thesynchronization signal S_(TRIG). The synchronization signal S_(TRIG) isgenerated by the ramp generator 70 and is supplied to the monitoringcircuit 80. The monitoring circuit 80 can detect the desired time windowfrom a defined logic level (e.g. a high level) of the signal S_(TRIG),for example (see top diagram in FIG. 8 ). The middle diagram in FIG. 8shows the frequency-divided oscillator signal s_(LO)′(t). The monitoringcircuit 80 is configured to count the cycles (e.g. on the basis of therising or falling edges) of the signal s_(LO)′(t) during the time windowT_(M). The average signal frequency during the time window T_(M) canthen be computed according to CNT/T_(M), CNT representing the count atthe end of the time window T_(M) (see bottom diagram in FIG. 8 ). Theaverage frequency f_(LO) can be inferred by multiplying the ratioCNT/T_(M) by the divisor value M (and possibly by the optional divisorvalue x, see FIG. 9 ), which is constant during operation.

FIG. 9 illustrates an example of a monitoring circuit 80 that isconfigured to implement the frequency measurement outlined in FIG. 8 .FIG. 9 also shows by way of example how an automatic test apparatus 2(ATE, automatic test equipment) can be connected to the MMIC 1 in whichthe monitoring circuit 80 is integrated, in order—for example as part ofa test at the end of the production line (end-of-line test, EOL test)—toread measured values for the frequency F_(LO). The ATE 2 may also beconfigured to take the measured values read for f_(LO) as a basis forcalibrating the VCO 60 (cf. FIG. 6 ), for example. In this regard, theparameter k_(VCO) can be adjusted by means of laser fusing of lines(e.g. strip lines) on the MMIC 10, for example. Effectively, this adaptsthe resonant frequency (center frequency) of the VCO for a givenactuating voltage.

According to the example depicted in FIG. 9 , the monitoring circuit 80comprises a counter 81 (count CNT), an evaluation unit 82, one or moreregisters 83 and a communication interface 84. In the present example,the counter 81 is activated and deactivated by means of thesynchronization signal S_(TRIG) provided by the ramp generator 70 (cf.FIG. 8 , bottom diagram). By way of example, the counter 81 counts theclock cycles of the signal s_(LO)′(t) only in that time window that isindicated by the signal S_(TRIG). In this regard, the counter 81 cane.g. be reset and activated on a rising edge of the signal S_(TRIG) anddeactivated on a falling edge of the signal S_(TRIG). The counter 81 canhave either the signal s_(LO)′(t) at the frequency f_(LO)′ supplied toit (output signal of the divider 72, cf. FIG. 5 ) or an againfrequency-divided signal at a frequency f_(LO)′/x (optional frequencydivider 85).

After completion of the measurement (see FIG. 8 , time t₁) at the end ofthe time window, the count CNT can be written to a register 83. At thesame time, the count CNT is checked by the evaluation unit 82, itessentially being detected whether the count CNT is in an admissibledesired range. Depending on the result of the check, the evaluation unit82 can write a Boolean output value to a register 83 (e.g. 0/fail,1/pass). The evaluation unit 82 can be implemented completely inhardware or (in part or in full) as software that is executed by aprocessor. In the case of a software implementation, the software can beexecuted in the system controller 50 (microcontroller, see FIG. 3 ), forexample. The register 83 may also be arranged in the system controller50. The ATE 2 can read the values in the registers 83 via thecommunication interface 84. In this regard, the ATE 2 can make contactwith the MMIC 10 to be tested (DUT, device under test), for example byone or more pins or test pads 8. The communication interface 84 maylikewise be part of the system controller 50 and may be configured totransmit digital data serially or in parallel. In an exemplaryembodiment, the communication interface 84 is an interface based on theSPI (serial peripheral interface) standard.

FIG. 10 illustrates an example of a method for testing or monitoring avoltage controlled RF oscillator of a local oscillator circuit (LOcircuit), such as e.g. the VCO 60 from FIG. 5 or 6 , which can be usedin the RF front end of a radar sensor. In the example depicted, the VCOis actuated such that it generates an RF oscillator signal s_(LO)(t) ata first frequency f_(LO) (see FIG. 10 , step S1). The first frequencyf_(LO) may be (adjustably) constant or can vary (e.g. at linearly risingor falling frequency). This actuating of the VCO can be performed indifferent ways. By way of example, the control voltage V_(CTRL) can beset directly (e.g. by means of a digital-to-analog convertor, forexample using the DAC 73 shown in FIG. 6 or another DAC), so that thephase locked loop (see FIG. 5 or 6 ) is not active (open-loop operationof the VCO). In an exemplary embodiment, the control voltage V_(CTRL)can also be generated directly by an automatic test apparatus (ATE,automatic test equipment). Alternatively, the ramp generator 70 andactive phase locked loop (closed-loop mode of the VCO) can be used togenerate a frequency profile (signal s_(RMP)(t)), for example afrequency ramp as depicted in FIG. 7 .

The frequency f_(LO) of the RF oscillator signal s_(LO)(t) is reduced toa frequency f_(LO)′ (second frequency) by means of a frequency divider.That is to say that the frequency divider takes the first RF oscillatorsignal s_(LO)(t) and generates an oscillator signal s_(LO)′(t) at thesecond frequency f_(LO)′, the frequency divider operating at an(adjustably) constant division ratio 1/M (see FIG. 10 , step S2). Thedivisor M is an integer (e.g. M=32), and the relationship f_(LO)=Mf_(LO)′ applies. The frequency f_(LO)′ of the oscillator signals_(LO)′(t) at the output of the frequency divider is measured and, as aresult of the measurement, at least one digital value is provided thatis dependent on the second frequency f_(LO)′ of the oscillator signals_(LO)′(t) (see FIG. 10 , step S3). The at least one digital value can,as explained with reference to FIGS. 8 and 9 , comprise a counter value(cf. FIG. 9 , counter 81) that represents the second frequency f_(LO)′(and hence also the first frequency f_(LO)). Alternatively oradditionally, the at least one digital value can comprise a Booleanvalue indicating whether the second frequency f_(LO)′—and hence also thefirst frequency f_(LO)—is in a desired range that is required. The atleast one digital value can be provided on a test contact (e.g. test pinor test pad, see FIG. 9 ) (FIG. 10 , step S4).

Providing a measured value representing the first frequency f_(LO) as adigital word (e.g. as a serial data stream) on a test contact allowscomparatively simple calibration of VCO parameter k_(VCO) by means of anATE (see FIG. 9 , ATE 2). In this case, the VCO is actuated at a definedvoltage V_(CTRL) and the resulting frequency f_(LO) is measured. If themeasured frequency differs from a desired value f_(LO,desired) by morethan a permitted difference Δf (f_(LO,desired)=V_(CTRL)·k_(VCO)), thenthe ATE can be used to adjust (tune) the parameter k_(VCO) of therelevant MMIC. The aforementioned desired range for the frequency f_(LO)is the range [f_(LO,desired)−Δf, f_(LO,desired)+Δf] in this case. FIG.11 illustrates a method for automatic tuning of the VCO parameterk_(VCO) performed by an ATE by way of example. In this case, the digitalvalue representing the frequency f_(LO) (or f_(LO)′) is read from theMMIC 10 by the ATE 2 digitally (FIG. 10 , step S5), and a differencebetween the frequency f_(LO) measured in the MMIC and an associateddesired value f_(LO,desired) is ascertained (FIG. 10 , step S6). Basedon the ascertained difference, the parameter k_(VCO) is fine-tuned (FIG.10 , step S7). The parameter k_(VCO) can be tuned by adapting the lengthof one or more strip lines (e.g. stubs) on the MMIC 10, for example.This allows the center frequency of the VCO (and hence also theparameter k_(VCO)) to be tuned. Suitable techniques for this, such ase.g. fusing of strip lines, are known per se and not explained furthernow.

The monitor circuit shown in FIG. 9 can be used not only for thefrequency measurement and the communication with an ATE, but also forself-tests during operation or during a self-test phase when the sensoris switched on. In this case, the check to determine whether the numberof clock cycles counted by the counter 82 matches a desired (average)frequency is performed by the evaluation unit 82, which delivers aBoolean value (e.g. “pass” (0) or “fail” (1)) as the result. Theevaluation unit 82 may, as mentioned, be embodied at least in part insoftware, which is executed on the system controller 50, for example. Ifthe evaluation unit 82 indicates an erroneous frequency value (“fail”),the system controller 50 can generate an error signal that signals theerror e.g. to a superordinate control unit.

The frequency measurement performed by the monitor unit 80 using thecounter 81 does not necessarily have to be performed at a constantfrequency f_(LO) of the VCO, but rather measurements are alsofacilitated while the frequency f_(LO) is changed, for example during achirp pulse (frequency ramp). If multiple measurements are performede.g. during the rise (ramp up) or fall (ramp down) of the frequencyf_(LO), the evaluation unit 82 can also assess the linearity of afrequency ramp. This instance of application is depicted in the timingdiagrams shown in FIG. 12 . The top timing diagram for FIG. 12 shows afrequency ramp as in FIG. 7 , the middle timing diagram shows thesynchronization signal S_(TRIG) provided by the ramp generator 70 andthe bottom timing diagram shows the (reduced-frequency) oscillatorsignal s_(LO)′(t). The time window within which the counter 81 countsthe cycles of the oscillator signal s_(LO)′(t) begins at the time t₂ andends at the time t₃.

In the example depicted in FIG. 12 , the count of the counter 81 is readand evaluated not only at the end of the time window (time t₃) but alsoat one or more times within the time window t₂ to t₃. In the presentexample, the count CNT of the counter is evaluated at the times t_(M1),t_(M2), t_(M3) and t₃. For each time t_(M1), t_(M2), t_(M3) and t₃, thecount is compared with an associated reference value, allowing improvedassessment of the linearity of a frequency ramp. In the example fromFIG. 12 (top diagram), the ideally linear frequency ramp is depicted asa solid line and an example of a differing real frequency ramp isdepicted as a dash-dot line. It can be seen that the real frequency ramp(dash-dot line) does not substantially differ from the ideal linebetween the times t₂ and t_(M2). A difference first occurs in the secondportion of the time window between the times t_(M2) and t₃ (that is tosay at higher frequencies). If only a yes/no decision (“pass” or “fail”)is needed, however, an evaluation at the end of the time window (i.e.after the time t₃) may suffice.

What is claimed is:
 1. An apparatus, comprising: an RF oscillatorconfigured to generate an RF oscillator signal at a first frequency; afrequency divider having a division ratio that is fixed duringoperation, to which the RF oscillator signal is supplied and which isconfigured to provide a second oscillator signal at a second frequency;a monitor circuit, to which the second oscillator signal at the secondfrequency is supplied and which is configured to measure the secondfrequency and provide at least one digital value that is dependent onthe second frequency of the second oscillator signal; and a rampgenerating circuit configured to generate a ramp signal, wherein themonitor circuit is configured to receive a synchronization signal, thesynchronization signal synchronizing a measurement of the secondfrequency by the monitor circuit to the generation of the ramp signal.2. The apparatus according to claim 1, further comprising: asynchronization signal generating circuit configured to generate thesynchronization signal and provide the synchronization signal to themonitor circuit.
 3. The apparatus according to claim 1, wherein thefrequency divider is a fixed frequency divider and the division ratio isfixed at a constant value throughout a generation of at least one rampof the ramp signal.
 4. The apparatus according to claim 1, wherein: thesynchronization signal defines a measurement time interval thatcoincides with a portion of the ramp signal selected for monitoring,wherein the measurement time interval is defined by a first transitionedge of the synchronization signal and a second transition edge of thesynchronization signal, and the monitor circuit is configured to enablemeasuring of the second frequency in response to detecting the firsttransition edge of the synchronization signal and disable measuring ofthe second frequency in response to detecting the second transition edgeof the synchronization signal.
 5. The apparatus according to claim 4,wherein the portion of the ramp signal selected for monitoring is aspecific frequency ramp of the ramp signal.
 6. The apparatus accordingto claim 5, wherein the portion of the ramp signal selected formonitoring is defined by a minimum oscillator frequency and a maximumoscillator frequency of the specific frequency ramp.
 7. The apparatusaccording to claim 4, wherein the monitor circuit is configured tocalculate an average value of the second frequency measured during themeasurement time interval, and convert the average value of the secondfrequency into an average value of the first frequency.
 8. The apparatusaccording to claim 4, wherein the monitor circuit is configured to counta total number of clock cycles of the second oscillator signal duringthe measurement time interval, compare the total number of clock cyclesto a tolerance range to determine whether the total number of clockcycles are within or outside the tolerance range, and generate a digitalvalue based on whether the total number of clock cycles is within oroutside the tolerance range.
 9. The apparatus according to claim 8,wherein the monitor circuit includes a counter that is configured tocount the total number of clock cycles of the second oscillator signalduring the measurement time interval, wherein the counter is activatedin response to the first transition edge of the synchronization signaland deactivated in response to the second transition edge of thesynchronization signal.
 10. The apparatus according to claim 4, whereinthe monitor circuit is configured to count a total number of clockcycles of the second oscillator signal during the measurement timeinterval, and generate a digital value representative of the totalnumber of clock cycles of the second oscillator signal.
 11. Theapparatus according to claim 1, wherein: the RF oscillator is connectedin a phase locked loop, the ramp generating circuit is coupled to thephase locked loop such that the first frequency is adjusted according tothe ramp signal, and the frequency divider is connected in a feedbackloop of the phase locked loop.
 12. A monolithic microwave integratedcircuit (MMIC), comprising: an RF oscillator, integrated in asemiconductor chip; a ramp generating circuit, integrated in thesemiconductor chip, the ramp generating circuit configured to generate aramp signal to control a frequency modulation of the RF oscillator; anda monitor circuit integrated in the semiconductor chip, the monitorcircuit comprising a counter configured to receive an RF signal derivedfrom the RF oscillator, the counter configured to provide at least onedigital value that is dependent on a number of cycles of the RF signalduring a measurement time interval and to receive a synchronizationsignal to synchronize the measurement time interval with a ramp of theramp signal.
 13. The MMIC according to claim 12, further comprising: asynchronization signal generating circuit integrated in thesemiconductor chip, the synchronization signal generating circuitconfigured to generate the synchronization signal and provide thesynchronization signal to the monitor circuit.
 14. The MMIC according toclaim 12, wherein: the synchronization signal defines a measurement timeinterval that coincides with the ramp of the ramp signal selected formonitoring, wherein the measurement time interval is defined by a firsttransition edge of the synchronization signal and a second transitionedge of the synchronization signal, and the monitor circuit isconfigured to enable counting of the number of cycles in response todetecting the first transition edge of the synchronization signal anddisable counting of the number of cycles in response to detecting thesecond transition edge of the synchronization signal.
 15. The MMICaccording to claim 14, wherein the monitor circuit is configured tocompare the number of cycles to a range to determine whether the numberof cycles are within or outside the range, and generate a digital valuebased on whether the number of cycles is within or outside the range.16. The MMIC according to claim 12, wherein the counter is activated inresponse to a first transition edge of the synchronization signal anddeactivated in response to a second transition edge of thesynchronization signal.
 17. The MMIC according to claim 16, wherein thefirst transition edge and the second transition edge are synchronizedwith a start and an end of the ramp, respectively.
 18. The MMICaccording to claim 12, wherein the number of cycles is a total number ofclock cycles dependent on a frequency of the RF signal, and the monitorcircuit is configured to generate a digital value representative of thetotal number of clock cycles that occur during the measurement timeinterval.
 19. A method, comprising: actuating an RF oscillator;generating a ramp signal to control a frequency modulation of the RFoscillator; counting a number of cycles of an RF signal derived from theRF oscillator; generating at least one digital value that is dependenton the number of cycles of the RF signal counted during a measurementtime interval; and synchronizing the measurement time interval with aramp of the ramp signal based on a synchronization signal.